{"id":351,"date":"2020-03-04T23:25:13","date_gmt":"2020-03-04T20:25:13","guid":{"rendered":"https:\/\/tomorrow82.ru\/?page_id=351"},"modified":"2020-03-04T23:25:13","modified_gmt":"2020-03-04T20:25:13","slug":"the-cascode-diff-amp-input-stage","status":"publish","type":"page","link":"https:\/\/tomorrow82.ru\/?page_id=351","title":{"rendered":"The Cascode Diff-Amp Input Stage"},"content":{"rendered":"\n

The input stage to the amplifier is shown in Figure 1. R1 sets the \ninput resistance at 20 kohm. R2 and C1 form a 200 kHz input low-pass \nfilter to protect the amplifier from unwanted RF signals at the input. \nQ1 through Q4 form a complementary diff amp input stage. The input \nsignal is applied to the bases of Q1 and Q3 while the feedback signal is\n applied to the bases of Q2 and Q4. The diff amps subtract the feedback \nsignal from the input signal to generate error signals which drive the \nfollowing stages in the amplifier. The error signals are the collector \ncurrents in Q1 and Q3. In addition to being part of the input low-pass \nfilter, C1 improves the bandwidth of the feedback signal path through \nthe diff amps for improved phase margin in the loop-gain transfer \nfunction.<\/p>\n\n\n\n

\"Figure<\/td><\/tr>
Figure 1. Amplifier input stage.<\/td><\/tr><\/tbody><\/table><\/figure>\n\n\n\n

Q5 and Q6 form cascode common-base stages which conduct the diff amp \nerror currents to the load resistors R11 and R12. Q5 and Q6 also \nfunction to reduce the voltage across Q1 — Q4 by about 18 V to prevent \nvoltage breakdown in these transistors. Diodes D13 through D16 are 20 V \nZener diodes which regulate the voltages that set the bias currents in \nthe diff amps. Each zener diode is biased at a current of about 3.3 mA. \nAlthough a single zener diode could be used in place of two series \ndiodes, I have found that the error tolerance in the voltage is less \nwith the series connection of 2 diodes. C2 through C5 are ac decoupling \ncapacitors which ensure the bases of Q5 and Q6 are at ac signal ground. \nR15 and R16 set the tail bias current in each diff amp to about 3.25 mA.\n This current and resistors R7 through R10 set the differential \ntansconductance gain of the diff amps at about 1.6 mA\/V. The diff amps \nuse resistive tail current bias circuits. Not only do these generate \nless noise than active current sources, but they provide a smooth \namplifier turn-on that is free of thumps. The voltage gain of each diff \namp is approximately 2, i.e. 6 dB.<\/p>\n\n\n\n

R7 through R10 are emitter degeneration resistors which play an \nimportant part in the frequency compensation of the amplifier. Without \nthese resistors, the gain of the diff amps would increase to about 40, \ni.e. to 32 dB. If the theory can be believed, the distortion would \nsimultaneously decrease by a factor of 40. However, this would seriously\n degrade the stability of the amplifier from oscillations unless the \ndiff-amp bias currents are decreased, the compensation capacitors in the\n second stage are increased, or both so as to maintain the same \ngain-bandwidth product. Either would degrade the slew rate.<\/p>\n\n\n\n

R7 through R10 not only decrease the gain of the diff amps, but they \nalso improve its linearity and dynamic range. This is illustrated in \nFigure 2 which shows plots of the collector currents in Q1 and Q2 (or Q3\n and Q4) as a function of the differential input voltage with and \nwithout the emitter resistors. The current I<\/em>Q<\/em><\/sub>\n is the diff-amp bias current which is about 3.25 mA. The linear range \nis taken to be the region between the dots where the currents vary \nbetween 5% and 95% of the maximum value. Without the resistors, the diff\n amps leave the linear range when the differential input voltage exceeds\n about 57 mV. With the emitter resistors, this voltage is increased to \nabout 951 mV, or by a factor of about 17 (24 dB). This reduces the \nsusceptibility of the amplifier to the transient types of distortion \nknown as slewing induced distortion (SID), transient intermodulation \ndistortion (TIM), and dynamic intermodulation distortion (DIM). The \nfigure shows that with the emitter resistors, the diff amp is linear \nwith input voltages as large as 1 V.<\/p>\n\n\n\n

\"Figure<\/td><\/tr>
Figure 2. Plots of the diff amp currents versus differential input voltage.<\/td><\/tr><\/tbody><\/table><\/figure>\n\n\n\n

Most of the amplifier circuits that I have seen either do not have \nemitter degeneration resistors in the input diff amps or the value of \nthe resistors is small compared to the 300 ohm values in the Low TIM \namplifier. I have even seen some circuits in which the resistors are \npresent but capacitors are used to bypass them for AC signals. These \namplifiers usually have a very high open-loop gain, i.e. the gain before\n feedback. This makes the distortion lower when feedback is added, but \nthey tend to oscillate unless large value compensation capacitors are \nused. This reduces the amplifier slew rate and increases the \nsusceptibility to transient distortion problems. <\/p>\n\n\n\n

The Low TIM amplifier is designed for a specified gain-bandwidth \nproduct and slew rate. I chose a gain-bandwidth product of about 8.5 \nMHz. This is low enough to qualify the amplifier as a low-feedback \ndesign. However, it is high enough to give both low distortion figures \nand a closed-loop bandwidth that is over 400 kHz before the addition of \nthe input low-pass filter (R2 and C1). I chose a slew rate of about 60 \nvolts per microsecond. This is much higher than required. It is high \nenough to give a large-signal bandwidth of around 220 kHz. However, I \nnever recommend anyone to test the amplifier at full power above 20 kHz.\n The output transistors are put under severe thermal stress during such \ntests, and they can fail. I know someone who used a Low TIM amplifier to\n drive an ultrasonic shaker table at full power at a frequency of 40 \nkHz. He said it worked, but I don’t recommend such applications.<\/p>\n\n\n\n

The bandwidth of the input low-pass filter (R2 and C1) is chosen so \nthat it is impossible for the amplifier to slew before it clips. Figure 3\n shows a plot of the differential input voltage v<\/em>ID<\/em><\/sub> to the diff amps for a step input voltage which drives the amplifier to the verge of clipping. The peak value occurs at time t<\/em>1<\/sub>\n which is about 0.6 microseconds. The peak value is 0.437 V. For a \nsquare wave input signal, this peak is increased by a factor of 2 to \n0.874 V, which is within the range of linear response of the diff amp \nillustrated in Figure 2. Thus the amplifier cannot slew before it clips \nwith a square-wave input signal. This is the worst case test for \nslewing.<\/p>\n\n\n\n

\"Figure<\/td><\/tr>
Figure 3. Plot of the differential input voltage for a voltage step input.<\/td><\/tr><\/tbody><\/table><\/figure>\n","protected":false},"excerpt":{"rendered":"

The input stage to the amplifier is shown in Figure 1. R1 sets the input resistance at 20 kohm. R2 and C1 form a 200 kHz input low-pass filter to protect the amplifier from unwanted RF signals at the input. Q1 through Q4 form a complementary diff amp input stage. The input signal is applied… <\/p>\n

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